In general, “fading” arises in radio communication, and causes degraded transmission quality, that is, significant degradation of the bit error rate characteristic.
As one method for compensating for degraded transmission quality caused by the phasing, “transmit diversity” is commonly known. In the following, one type of transmit diversity, “closed loop type transmit diversity mode 1”, is described (see non-patent document 1).
FIG. 1 illustrates an exemplary configuration of a transmission unit of a closed loop type transmit diversity, and FIG. 2 illustrates an exemplary configuration of a reception unit of the closed loop type transmit diversity. In a closed loop type transmit diversity, two sequences of transmitted data are multiplied with respective complex weights based on a feedback information (FBI) bit from a mobile station,w1=A1eiφ1;andw2=A2eiφ2.Then, a spreading operation is performed on them, and the resulting sequences are transmitted. First, a radio base station transmits common pilot channels (CPICH) with the same carrier phase via two antennas. The CPICHs transmitted via the two antennas spread with the same spreading code, and orthogonalization is achieved by changing a pilot symbol. A mobile station generates a FBI bit for controlling a reception carrier phase difference of a signal separated after inverse spreading of the CPICHs transmitted via the two antennas in a reception device thereof, and transmits the FBI bit over a dedicated physical control channel (DPCCH) in an uplink dedicated physical channel (DPCH). By controlling the transmission carrier phase of antenna 2 with use of the FBI bit from the mobile station, it is possible to reduce bit errors caused by reception signal power decrease due to fading. In a transmission unit of a radio base station, antenna weights w1 and w2 generated based on the FBI bit from the mobile station are multiplied with sequences of transmitted data for two antennas, and the resulting sequences are transmitted.
A closed loop type diversity mode 1, which is specified in accordance with 3GPP, controls the transmission carrier phase of DPCH of the second antenna with resolution of π/4 carrier phase so that received signals via the two antennas can have approximately the same phase in reception at the mobile station. In the following, an exemplary operation in the case where closed loop type transmit diversity mode 1 is applied to a dedicated physical channel DPCH is described in detail.
Transmission amplitude of two antennas in the slot n is represented as
            A              1        ,        n              =                  A                  2          ,          n                    =              1                  2                      ,Transmission carrier phase is represented asφ1,n=0, and φ2,n={±π/4,±3π/4}.In other words, the weights w1 and w2 have values
                    w        1            =              1                  2                      ,    and                      w        2            =                        1                      2                          ⁢                  ⅇ                      1                          ϕ                              2                ,                n                                                          ,  respectively.
The mobile station computes w under which P is maximum in the following formula,P=wHHHHw  (1),where H=[h1, h2] and w=[w1, w2]T, and h1 and h2 are as follows.
h1: estimated channel impulse response from transmission antenna 1 (derived from CPICH from the transmission antenna 1)
h2: estimated channel impulse response from transmission antenna 2 (derived from CPICH from the transmission antenna 2)
Specifically, the mobile station estimates reception carrier phases θ1,nCP and θ2,nCP transmitted by the two antennas, and generates FBI bit bn in the slot n. Here, assuming that
            w      2        =                  1                  2                    ⁢              ⅇ                  ⅈϕ                      2            ,            n                                ,the following equation holdsφ2,n=θ1,nCP−θ2,nCP.Since Φ2,n has a discrete value, it is computed as follows.
For an even slot n, if −π/2≦(θ1,nCP−θ2,nCP)≦π, then bn=0, otherwise bn=1. For an odd slot n, if 0≦(θ1,nCP-θ2,nCP)≦π, then bn=0, otherwise bn=1.
The radio base station determines a temporal transmission carrier phase φ2,(n+1) in the (n+1)th slot of DPCH in the second antenna depending on a decode result {circumflex over (b)}n of the FBI bit (if no FBI bit exists, {circumflex over (b)}n=bn) as follows.
For an even number n, if {circumflex over (b)}n=0, then φ2,(n+1)=0, otherwise φ2,(n+1)=π. For an odd number n, if {circumflex over (b)}n=0, then φ2,(n+1)=π/2, otherwise φ2,(n+1)=−π/2. Finally, the transmission carrier phase Φ2,(n+1) of the second antenna in the (n+1)th slot is derived from temporal carrier phases of the slots n and (n+1) as follows;φ2,(n+1)=(φ2,n+φ2,(n+1))/2.
Meanwhile, in a commonly used mobile communication system, an area covered by a single radio base station is partitioned into a plurality of smaller cells (also referred to as sectors), and communications are actually carried out for each cell. FIG. 3 is a schematic view illustrating exemplary cells in the mobile communication system. In FIG. 3, an exemplary configuration where three radio base stations 1-3 each includes three respective sectors is illustrated. Communication control for communicating to a plurality of cells simultaneously, which is referred to as soft hand over, is carried out in a dedicated physical channel (DPCH) in a 3GPP downlink. For the simultaneously communicating cells, there are two cases. One is the case where the cells are different cells within the same base station. The other is the case where the cells are cells within different base stations. In these cases, the simultaneous communications within different cells within the same base station are called intra-cell handover, and the simultaneous communications within plural cells belonging to different base stations are called inter-cell handover. In FIG. 3, a mobile station 1 performs the intra-cell handover between cells 11 and 12 within a radio base station 1, whereas a mobile station 2 performs the inter-cell handover between a cell 23 in a radio base station and a cell 31 in a radio base station 3.
Then, if the closed loop type transmit diversity control is carried out and the soft handover is performed, the mobile station generates a FBI bit for signals from all the simultaneously communicating cells so that reception carrier phase difference can be optimal. One specific computation of the FBI bit is as follows.
In the case of the soft handover (in the case of communications to plural cells), the vector w is derived so that P as presented below is maximized.P=wH(H1HH1+H2HH2+ . . . )w  (2),where Hi is an estimated channel impulse response associated with the ith cell.
In the above description, the operation corresponding to the case where transmit diversity is applied to a downlink dedicated physical channel DPCH in soft handover has been described. Next, an exemplary operation corresponding to the case where closed loop type transmit diversity mode 1 is applied to a high speed physical downlink shared channel (HS-PDSCH) being a downlink shared channel is described below.
HS-PDSCH is a shared physical channel for carrying data in transmission scheme HSDPA (High Speed Downlink Packet Access) for fast downlink data transmission. Other physical channel include HS-SCCH (High Speed Shared Control Channel) being a shared control channel and A-DPCH (Associated-Dedicated Physical Channel) dedicated to each mobile station.
Similar to the case of a dedicated physical channel, the transmit diversity in HSDPA generates a FBI bit from the phase difference of CPICHs from the two antennas, transmits the FBI bit in an uplink DPCH, and controls the phase of HS-PDSCH from the second antenna in a downlink. However, HS-PDSCH in HSDPA does not perform the soft handover like the dedicated physical channel (i.e. does not communicate to multiple cells simultaneously), but always communities to a single cell. In other words, the vector w computed with the formula (1) becomes the optimum.
In HSDPA, HS-PDSCH communicates to a single cell, whereas the other dedicated physical channels communicate to a plurality of cells. Thus, since the optimum vector w is computed with the respective formulae (1) and (2), it is difficult to compute the optimum vector w for both of them. Thus, the non-patent document 2 discloses that the vector w is derived with the following formula.P=wH(α(H1HH1)+(1−α)(H2HH2+H3HH3+ . . . )w  (3),where H1 represents an estimated channel impulse response associated with a cell communicating by HS-PDSCH, H2, H3, . . . estimated channel impulse responses associated with cells communicating not by HS-PDSCH but by dedicated physical channels, and α is a real value ranging between 0 and 1.
For example, if α is set as 0.7, it is possible to compute a more appropriate w for not only HS-PDSCH but also the dedicated physical channels.
Also, the non-patent document 2 discloses some technique referred to as Fast Adaptive Emphasis. Specifically, in the case where the above-mentioned HD-PDSCH only communicates within a single cell and the dedicated physical channel communicates with a plurality of cells, the parameter α has different values depending on whether HS-PDSCH is assigned to the relevant mobile station. By using this technique, only if HS-PDSCH is transmitted in burst, the optimal α to HS-PDSCH can be used to compute antenna weights without transmission of the HS-PDSCH, whereas when only the dedicated physical channel is transmitted, the optimal α to the dedicated physical channel can be used to compute the antenna weights.
Non-patent document 1: 3GPP TS25.214 v5.a.0
Non-patent document 2: 3GPP TR25.899 v6.1.0